Asymmetrical MIMO wireless communications

ABSTRACT

A method for asymmetrical MIMO wireless communication begins by determining a number of transmission antennas for the asymmetrical MIMO wireless communication. The method continues by determining a number of reception antennas for the asymmetrical MIMO wireless communication. The method continues by, when the number of transmission antennas exceeds the number of reception antennas, using spatial time block coding for the asymmetrical MIMO wireless communication. The method continues by, when the number of transmission antennas does not exceed the number of reception antennas, using spatial multiplexing for the asymmetrical MIMO wireless communication.

CROSS REFERENCE TO RELATED APPLICATIONS

This invention is claiming priority under 35 USC §119 as a continuationof the patent application entitled, ASYMMETRICAL MIMO WIRELESSCOMMUNICATIONS, having Ser. No. 10/979,368, and filed on Nov. 1, 2004,that itself claims priority under 35 USC §119(e) to three co-pendingprovisional patent applications: the first is entitled WLAN TRANSMITTERHAVING HIGH DATA THROUGHPUT having a filing date of Feb. 19, 2004 and aprovisional Ser. No. 60/545,854; the second is entitled MULTIPLE INPUTMULTIPLE OUTPUT WIRELESS LOCAL AREA NETWORK COMMUNICATIONS having aprovisional Ser. No. 60/556,264 and a filing date of Mar. 25, 2004; andthe third has the same title as the present patent application, aprovisional Ser. No. 60/575,920, and a provisional filing date of Jun.1, 2004.

BACKGROUND OF THE INVENTION

1. Technical Field of the Invention

This invention relates generally to wireless communication systems andmore particularly to a transmitter transmitting at high data rateswithin such wireless communication systems.

2. Description of Related Art

Communication systems are known to support wireless and wire linedcommunications between wireless and/or wire lined communication devices.Such communication systems range from national and/or internationalcellular telephone systems to the Internet to point-to-point in-homewireless networks. Each type of communication system is constructed, andhence operates, in accordance with one or more communication standards.For instance, wireless communication systems may operate in accordancewith one or more standards including, but not limited to, IEEE 802.11,Bluetooth, advanced mobile phone services (AMPS), digital AMPS, globalsystem for mobile communications (GSM), code division multiple access(CDMA), local multi-point distribution systems (LMDS),multi-channel-multi-point distribution systems (MMDS), and/or variationsthereof.

Depending on the type of wireless communication system, a wirelesscommunication device, such as a cellular telephone, two-way radio,personal digital assistant (PDA), personal computer (PC), laptopcomputer, home entertainment equipment, et cetera communicates directlyor indirectly with other wireless communication devices. For directcommunications (also known as point-to-point communications), theparticipating wireless communication devices tune their receivers andtransmitters to the same channel or channels (e.g., one of the pluralityof radio frequency (RF) carriers of the wireless communication system)and communicate over that channel(s). For indirect wirelesscommunications, each wireless communication device communicates directlywith an associated base station (e.g., for cellular services) and/or anassociated access point (e.g., for an in-home or in-building wirelessnetwork) via an assigned channel. To complete a communication connectionbetween the wireless communication devices, the associated base stationsand/or associated access points communicate with each other directly,via a system controller, via the public switch telephone network, viathe Internet, and/or via some other wide area network.

For each wireless communication device to participate in wirelesscommunications, it includes a built-in radio transceiver (i.e., receiverand transmitter) or is coupled to an associated radio transceiver (e.g.,a station for in-home and/or in-building wireless communicationnetworks, RF modem, etc.). As is known, the receiver is coupled to theantenna and includes a low noise amplifier, one or more intermediatefrequency stages, a filtering stage, and a data recovery stage. The lownoise amplifier receives inbound RF signals via the antenna andamplifies then. The one or more intermediate frequency stages mix theamplified RF signals with one or more local oscillations to convert theamplified RF signal into baseband signals or intermediate frequency (IF)signals. The filtering stage filters the baseband signals or the IFsignals to attenuate unwanted out of band signals to produce filteredsignals. The data recovery stage recovers raw data from the filteredsignals in accordance with the particular wireless communicationstandard.

As is also known, the transmitter includes a data modulation stage, oneor more intermediate frequency stages, and a power amplifier. The datamodulation stage converts raw data into baseband signals in accordancewith a particular wireless communication standard. The one or moreintermediate frequency stages mix the baseband signals with one or morelocal oscillations to produce RF signals. The power amplifier amplifiesthe RF signals prior to transmission via an antenna.

Typically, the transmitter will include one antenna for transmitting theRF signals, which are received by a single antenna, or multipleantennas, of a receiver. When the receiver includes two or moreantennas, the receiver will select one of them to receive the incomingRF signals. In this instance, the wireless communication between thetransmitter and receiver is a single-output-single-input (SISO)communication, even if the receiver includes multiple antennas that areused as diversity antennas (i.e., selecting one of them to receive theincoming RF signals). For SISO wireless communications, a transceiverincludes one transmitter and one receiver. Currently, most wirelesslocal area networks (WLAN) that are IEEE 802.11, 802.11a, 802,11b, or802.11g employ SISO wireless communications.

Other types of wireless communications includesingle-input-multiple-output (SIMO), multiple-input-single-output(MISO), and multiple-input-multiple-output (MIMO). In a SIMO wirelesscommunication, a single transmitter processes data into radio frequencysignals that are transmitted to a receiver. The receiver includes two ormore antennas and two or more receiver paths. Each of the antennasreceives the RF signals and provides them to a corresponding receiverpath (e.g., LNA, down conversion module, filters, and ADCs). Each of thereceiver paths processes the received RF signals to produce digitalsignals, which are combined and then processed to recapture thetransmitted data.

For a multiple-input-single-output (MISO) wireless communication, thetransmitter includes two or more transmission paths (e.g., digital toanalog converter, filters, up-conversion module, and a power amplifier)that each converts a corresponding portion of baseband signals into RFsignals, which are transmitted via corresponding antennas to a receiver.The receiver includes a single receiver path that receives the multipleRF signals from the transmitter. In this instance, the receiver usesbeam forming to combine the multiple RF signals into one signal forprocessing.

For a multiple-input-multiple-output (MIMO) wireless communication, thetransmitter and receiver each include multiple paths. In such acommunication, the transmitter parallel processes data using a spatialand time encoding function to produce two or more streams of data. Thetransmitter includes multiple transmission paths to convert each streamof data into multiple RF signals. The receiver receives the multiple RFsignals via multiple receiver paths that recapture the streams of datautilizing a spatial and time decoding function. The recaptured streamsof data are combined and subsequently processed to recover the originaldata.

With the various types of wireless communications (e.g., SISO, MISO,SIMO, and MIMO), it would be desirable to use one or more types ofwireless communications to enhance data throughput within a WLAN. Forexample, high data rates can be achieved with MIMO communications incomparison to SISO communications. However, most WLAN include legacywireless communication devices (i.e., devices that are compliant with anolder version of a wireless communication standard. As such, atransmitter capable of MIMO wireless communications should also bebackward compatible with legacy devices to function in a majority ofexisting WLANs.

Further, asymmetric MIMO is any M transmit antenna, N receive antennacommunication system in which M!=N. The case of M<N does not need to beconsidered by the 802.11n standard. However, the practical case of M>Ndoes. This case may happen when an access point (AP) with many antennasattempts to send frames to devices with only two (e.g. an old laptopPC). This case may also be relevant to certain broadcast/multicastGreenfield frame transmissions, e.g. beacons.

Therefore, a need exists for a WLAN transmitter that is capable of highdata throughput, is backward compatible with legacy devices, andsupports asymmetrical MIMO transmissions.

BRIEF SUMMARY OF THE INVENTION

The asymmetrical wireless communication of the present inventionsubstantially meets these needs and others. In one embodiment, a methodfor asymmetrical MIMO wireless communication begins by determining anumber of transmission antennas for the asymmetrical MIMO wirelesscommunication. The method continues by determining whether theasymmetric MIMO wireless communication will use spatial multiplexing orspace-time block coding based on the number of transmission antennas.The method continues, for spatial multiplexing, by providing differentconstellation points for transmission on each of the number oftransmission antennas. The method continues by, for space-time blockcoding: during a first time interval: providing, for transmission on afirst one of the number of transmission antennas, a first constellationpoint, and providing, for transmission on a second one of the number oftransmission antennas, a second constellation point. The methodcontinues by, during a second time interval: providing, for transmissionon the first one of the number of transmission antennas, an inversecomplex conjugate of the second constellation point, and providing, fortransmission on the second one of the number of transmission antennas, acomplex conjugate of the second constellation point.

In another embodiment, a method for asymmetrical MIMO wirelesscommunication begins by determining a number of transmission antennasfor the asymmetrical MIMO wireless communication. The method continuesby determining a number of reception antennas for the asymmetrical MIMOwireless communication. The method continues by, when the number oftransmission antennas exceeds the number of reception antennas, usingspatial time block coding for the asymmetrical MIMO wirelesscommunication. The method continues by, when the number of transmissionantennas does not exceed the number of reception antennas, using spatialmultiplexing for the asymmetrical MIMO wireless communication.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

FIG. 1 is a schematic block diagram of a wireless communication systemin accordance with the present invention;

FIG. 2 is a schematic block diagram of a wireless communication devicein accordance with the present invention;

FIG. 3 is a schematic block diagram of an RF transmitter in accordancewith the present invention;

FIG. 4 is a schematic block diagram of an RF receiver in accordance withthe present invention;

FIG. 5 is a logic diagram of a method for baseband processing of data inaccordance with the present invention;

FIG. 6 is a logic diagram of a method that further defines Step 120 ofFIG. 5.

FIGS. 7-9 illustrate logic diagrams of various embodiments for encodingthe scrambled data in accordance with the present invention;

FIGS. 10A and 10B are a schematic block diagram of a radio transmitterin accordance with the present invention;

FIGS. 11A and 11B are a schematic block diagram of a radio receiver inaccordance with the present invention;

FIG. 12 is a schematic block diagram of a channel encoder in accordancewith the present invention;

FIG. 13 is a schematic block diagram of a constituent encoder inaccordance with the present invention;

FIG. 14 is a schematic block diagram of an alternate embodiment of aconstituent encoder in accordance with the present invention;

FIG. 15 is a schematic block diagram of a rate ⅖ encoder in accordancewith the present invention;

FIG. 16 is a schematic block diagram of a puncture encoder in accordancewith the present invention;

FIG. 17 is a schematic block diagram of another embodiment of a punctureencoder in accordance with the present invention;

FIG. 18 is a schematic block diagram of a low density parity checkencoder in accordance with the present invention;

FIG. 19 is an illustration of an interleaver in accordance with thepresent invention;

FIG. 20 is a diagram of an example asymmetric communication inaccordance with the present invention;

FIG. 21 is a diagram of space-time block coding in accordance with thepresent invention;

FIG. 22 is a diagram of two antenna asymmetrical wireless communicationin accordance with the present invention;

FIG. 23 is a diagram of three antenna asymmetrical wirelesscommunication in accordance with the present invention;

FIG. 24 is another diagram of three antenna asymmetrical wirelesscommunication in accordance with the present invention;

FIG. 25 is a diagram of four antenna asymmetrical wireless communicationin accordance with the present invention; and

FIG. 26 is another diagram of four antenna asymmetrical wirelesscommunication in accordance with the present invention;

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 is a schematic block diagram illustrating a communication system10 that includes a plurality of base stations and/or access points12-16, a plurality of wireless communication devices 18-32 and a networkhardware component 34. The wireless communication devices 18-32 may belaptop host computers 18 and 26, personal digital assistant hosts 20 and30, personal computer hosts 24 and 32 and/or cellular telephone hosts 22and 28. The details of the wireless communication devices will bedescribed in greater detail with reference to FIG. 2.

The base stations or access points 12-16 are operably coupled to thenetwork hardware 34 via local area network connections 36, 38 and 40.The network hardware 34, which may be a router, switch, bridge, modem,system controller, et cetera provides a wide area network connection 42for the communication system 10. Each of the base stations or accesspoints 12-16 has an associated antenna or antenna array to communicatewith the wireless communication devices in its area. Typically, thewireless communication devices register with a particular base stationor access point 12-14 to receive services from the communication system10. For direct connections (i.e., point-to-point communications),wireless communication devices communicate directly via an allocatedchannel.

Typically, base stations are used for cellular telephone systems andlike-type systems, while access points are used for in-home orin-building wireless networks. Regardless of the particular type ofcommunication system, each wireless communication device includes abuilt-in radio and/or is coupled to a radio. The radio includes a highlylinear amplifier and/or programmable multi-stage amplifier as disclosedherein to enhance performance, reduce costs, reduce size, and/or enhancebroadband applications.

FIG. 2 is a schematic block diagram illustrating a wirelesscommunication device that includes the host device 18-32 and anassociated radio 60. For cellular telephone hosts, the radio 60 is abuilt-in component. For personal digital assistants hosts, laptop hosts,and/or personal computer hosts, the radio 60 may be built-in or anexternally coupled component.

As illustrated, the host device 18-32 includes a processing module 50,memory 52, radio interface 54, input interface 58 and output interface56. The processing module 50 and memory 52 execute the correspondinginstructions that are typically done by the host device. For example,for a cellular telephone host device, the processing module 50 performsthe corresponding communication functions in accordance with aparticular cellular telephone standard.

The radio interface 54 allows data to be received from and sent to theradio 60. For data received from the radio 60 (e.g., inbound data), theradio interface 54 provides the data to the processing module 50 forfurther processing and/or routing to the output interface 56. The outputinterface 56 provides connectivity to an output display device such as adisplay, monitor, speakers, et cetera such that the received data may bedisplayed. The radio interface 54 also provides data from the processingmodule 50 to the radio 60. The processing module 50 may receive theoutbound data from an input device such as a keyboard, keypad,microphone, et cetera via the input interface 58 or generate the dataitself. For data received via the input interface 58, the processingmodule 50 may perform a corresponding host function on the data and/orroute it to the radio 60 via the radio interface 54.

Radio 60 includes a host interface 62, a baseband processing module 64,memory 66, a plurality of radio frequency (RF) transmitters 68-72, atransmit/receive (T/R) module 74, a plurality of antennas 82-86, aplurality of RF receivers 76-80, and a local oscillation module 100. Thebaseband processing module 64, in combination with operationalinstructions stored in memory 66, execute digital receiver functions anddigital transmitter functions, respectively. The digital receiverfunctions, as will be described in greater detail with reference to FIG.11B, include, but are not limited to, digital intermediate frequency tobaseband conversion, demodulation, constellation demapping, decoding,de-interleaving, fast Fourier transform, cyclic prefix removal, spaceand time decoding, and/or descrambling. The digital transmitterfunctions, as will be described in greater detail with reference toFIGS. 5-19, include, but are not limited to, scrambling, encoding,interleaving, constellation mapping, modulation, inverse fast Fouriertransform, cyclic prefix addition, space and time encoding, and/ordigital baseband to IF conversion. The baseband processing modules 64may be implemented using one or more processing devices. Such aprocessing device may be a microprocessor, micro-controller, digitalsignal processor, microcomputer, central processing unit, fieldprogrammable gate array, programmable logic device, state machine, logiccircuitry, analog circuitry, digital circuitry, and/or any device thatmanipulates signals (analog and/or digital) based on operationalinstructions. The memory 66 may be a single memory device or a pluralityof memory devices. Such a memory device may be a read-only memory,random access memory, volatile memory, non-volatile memory, staticmemory, dynamic memory, flash memory, and/or any device that storesdigital information. Note that when the processing module 64 implementsone or more of its functions via a state machine, analog circuitry,digital circuitry, and/or logic circuitry, the memory storing thecorresponding operational instructions is embedded with the circuitrycomprising the state machine, analog circuitry, digital circuitry,and/or logic circuitry.

In operation, the radio 60 receives outbound data 88 from the hostdevice via the host interface 62. The baseband processing module 64receives the outbound data 88 and, based on a mode selection signal 102,produces one or more outbound symbol streams 90. The mode selectionsignal 102 will indicate a particular mode as are illustrated in themode selection tables, which appear at the end of the detaileddiscussion. For example, the mode selection signal 102, with referenceto table 1 may indicate a frequency band of 2.4 GHz, a channel bandwidthof 20 or 22 MHz and a maximum bit rate of 54 megabits-per-second. Inthis general category, the mode selection signal will further indicate aparticular rate ranging from 1 megabit-per-second to 54megabits-per-second. In addition, the mode selection signal willindicate a particular type of modulation, which includes, but is notlimited to, Barker Code Modulation, BPSK, QPSK, CCK, 16 QAM and/or 64QAM. As is further illustrated in table 1, a code rate is supplied aswell as number of coded bits per subcarrier (NBPSC), coded bits per OFDMsymbol (NCBPS), data bits per OFDM symbol (NDBPS).

The mode selection signal may also indicate a particular channelizationfor the corresponding mode which for the information in table 1 isillustrated in table 2. As shown, table 2 includes a channel number andcorresponding center frequency. The mode select signal may furtherindicate a power spectral density mask value which for table 1 isillustrated in table 3. The mode select signal may alternativelyindicate rates within table 4 that has a 5 GHz frequency band, 20 MHzchannel bandwidth and a maximum bit rate of 54 megabits-per-second. Ifthis is the particular mode select, the channelization is illustrated intable 5. As a further alternative, the mode select signal 102 mayindicate a 2.4 GHz frequency band, 20 MHz channels and a maximum bitrate of 192 megabits-per-second as illustrated in table 6. In table 6, anumber of antennas may be utilized to achieve the higher bandwidths. Inthis instance, the mode select would further indicate the number ofantennas to be utilized. Table 7 illustrates the channelization for theset-up of table 6. Table 8 illustrates yet another mode option where thefrequency band is 2.4 GHz, the channel bandwidth is 20 MHz and themaximum bit rate is 192 megabits-per-second. The corresponding table 8includes various bit rates ranging from 12 megabits-per-second to 216megabits-per-second utilizing 2-4 antennas and a spatial time encodingrate as indicated. Table 9 illustrates the channelization for table 8.The mode select signal 102 may further indicate a particular operatingmode as illustrated in table 10, which corresponds to a 5 GHz frequencyband having 40 MHz frequency band having 40 MHz channels and a maximumbit rate of 486 megabits-per-second. As shown in table 10, the bit ratemay range from 13.5 megabits-per-second to 486 megabits-per-secondutilizing 1-4 antennas and a corresponding spatial time code rate. Table10 further illustrates a particular modulation scheme code rate andNBPSC values. Table 11 provides the power spectral density mask fortable 10 and table 12 provides the channelization for table 10.

The baseband processing module 64, based on the mode selection signal102 produces the one or more outbound symbol streams 90, as will befurther described with reference to FIGS. 5-9 from the output data 88.For example, if the mode selection signal 102 indicates that a singletransmit antenna is being utilized for the particular mode that has beenselected, the baseband processing module 64 will produce a singleoutbound symbol stream 90. Alternatively, if the mode select signalindicates 2, 3 or 4 antennas, the baseband processing module 64 willproduce 2, 3 or 4 outbound symbol streams 90 corresponding to the numberof antennas from the output data 88.

Depending on the number of outbound streams 90 produced by the basebandmodule 64, a corresponding number of the RF transmitters 68-72 will beenabled to convert the outbound symbol streams 90 into outbound RFsignals 92. The implementation of the RF transmitters 68-72 will befurther described with reference to FIG. 3. The transmit/receive module74 receives the outbound RF signals 92 and provides each outbound RFsignal to a corresponding antenna 82-86.

When the radio 60 is in the receive mode, the transmit/receive module 74receives one or more inbound RF signals via the antennas 82-86. The T/Rmodule 74 provides the inbound RF signals 94 to one or more RF receivers76-80. The RF receiver 76-80, which will be described in greater detailwith reference to FIG. 4, converts the inbound RF signals 94 into acorresponding number of inbound symbol streams 96. The number of inboundsymbol streams 96 will correspond to the particular mode in which thedata was received (recall that the mode may be any one of the modesillustrated in tables 1-12). The baseband processing module 60 receivesthe inbound symbol streams 90 and converts them into inbound data 98,which is provided to the host device 18-32 via the host interface 62.

As one of average skill in the art will appreciate, the wirelesscommunication device of FIG. 2 may be implemented using one or moreintegrated circuits. For example, the host device may be implemented onone integrated circuit, the baseband processing module 64 and memory 66may be implemented on a second integrated circuit, and the remainingcomponents of the radio 60, less the antennas 82-86, may be implementedon a third integrated circuit. As an alternate example, the radio 60 maybe implemented on a single integrated circuit. As yet another example,the processing module 50 of the host device and the baseband processingmodule 64 may be a common processing device implemented on a singleintegrated circuit. Further, the memory 52 and memory 66 may beimplemented on a single integrated circuit and/or on the same integratedcircuit as the common processing modules of processing module 50 and thebaseband processing module 64.

FIG. 3 is a schematic block diagram of an embodiment of an RFtransmitter 68-72. The RF transmitter 68-72 includes a digital filterand up-sampling module 75, a digital-to-analog conversion module 77, ananalog filter 79, and up-conversion module 81, a power amplifier 83 anda RF filter 85. The digital filter and up-sampling module 75 receivesone of the outbound symbol streams 90 and digitally filters it and thenup-samples the rate of the symbol streams to a desired rate to producethe filtered symbol streams 87. The digital-to-analog conversion module77 converts the filtered symbols 87 into analog signals 89. The analogsignals may include an in-phase component and a quadrature component.

The analog filter 79 filters the analog signals 89 to produce filteredanalog signals 91. The up-conversion module 81, which may include a pairof mixers and a filter, mixes the filtered analog signals 91 with alocal oscillation 93, which is produced by local oscillation module 100,to produce high frequency signals 95. The frequency of the highfrequency signals 95 corresponds to the frequency of the RF signals 92.

The power amplifier 83 amplifies the high frequency signals 95 toproduce amplified high frequency signals 97. The RF filter 85, which maybe a high frequency band-pass filter, filters the amplified highfrequency signals 97 to produce the desired output RF signals 92.

As one of average skill in the art will appreciate, each of the radiofrequency transmitters 68-72 will include a similar architecture asillustrated in FIG. 3 and further include a shut-down mechanism suchthat when the particular radio frequency transmitter is not required, itis disabled in such a manner that it does not produce interferingsignals and/or noise.

FIG. 4 is a schematic block diagram of each of the RF receivers 76-80.In this embodiment, each of the RF receivers 76-80 includes an RF filter101, a low noise amplifier (LNA) 103, a programmable gain amplifier(PGA) 105, a down-conversion module 107, an analog filter 109, ananalog-to-digital conversion module 111 and a digital filter anddown-sampling module 113. The RF filter 101, which may be a highfrequency band-pass filter, receives the inbound RF signals 94 andfilters them to produce filtered inbound RF signals. The low noiseamplifier 103 amplifies the filtered inbound RF signals 94 based on again setting 115 and provides the amplified signals to the programmablegain amplifier 105. The programmable gain amplifier further amplifiesthe inbound RF signals 94 before providing them to the down-conversionmodule 107.

The down-conversion module 107 includes a pair of mixers, a summationmodule, and a filter to mix the inbound RF signals with a localoscillation (LO) that is provided by the local oscillation module toproduce analog baseband signals. The analog filter 109 filters theanalog baseband signals and provides them to the analog-to-digitalconversion module 111 which converts them into a digital signal. Thedigital filter and down-sampling module 113 filters the digital signalsand then adjusts the sampling rate to produce the inbound symbol stream96.

FIG. 5 is a logic diagram of a method for converting outbound data 88into one or more outbound symbol streams 90 by the baseband processingmodule 64. The process begins at Step 110 where the baseband processingmodule receives the outbound data 88 and a mode selection signal 102.The mode selection signal may indicate any one of the various modes ofoperation as indicated in tables 1-12. The process then proceeds to Step112 where the baseband processing module scrambles the data inaccordance with a pseudo random sequence to produce scrambled data. Notethat the pseudo random sequence may be generated from a feedback shiftregister with the generator polynomial of S(x)=x⁷+x⁴+1.

The process then proceeds to Step 114 where the baseband processingmodule selects one of a plurality of encoding modes based on the modeselection signal. The process then proceeds to Step 116 where thebaseband processing module encodes the scrambled data in accordance witha selected encoding mode to produce encoded data. The encoding may bedone utilizing a parallel concatenated turbo encoding scheme and/or alow density parity check block encoding scheme. Such encoding schemeswill be described in greater detail with reference to FIGS. 12-19.Alternatively, the encoding may be done as further described in FIGS.7-9 which will be described below.

The process then proceeds to Step 118 where the baseband processingmodule determines a number of transmit streams based on the mode selectsignal. For example, the mode select signal will select a particularmode which indicates that 1, 2, 3, 4 or more antennas may be utilizedfor the transmission. Accordingly, the number of transmit streams willcorrespond to the number of antennas indicated by the mode selectsignal. The process then proceeds to Step 120 where the basebandprocessing module converts the encoded data into streams of symbols inaccordance with the number of transmit streams in the mode selectsignal. This step will be described in greater detail with reference toFIG. 6.

FIG. 6 is a logic diagram of a method performed by the basebandprocessing module to convert the encoded data into streams of symbols inaccordance with the number of transmit streams and the mode selectsignal. Such processing begins at Step 122 where the baseband processingmodule interleaves the encoded data over multiple symbols andsubcarriers of a channel to produce interleaved data. In general, theinterleaving process is designed to spread the encoded data overmultiple symbols and transmit streams. This allows improved detectionand error correction capability at the receiver. In one embodiment, theinterleaving process will follow the IEEE 802.11(a) or (g) standard forbackward compatible modes. For higher performance modes (e.g., IEEE802.11(n), the interleaving will also be done over multiple transmitpaths or streams.

The process then proceeds to Step 124 where the baseband processingmodule demultiplexes the interleaved data into a number of parallelstreams of interleaved data. The number of parallel streams correspondsto the number of transmit streams, which in turn corresponds to thenumber of antennas indicated by the particular mode being utilized. Theprocess then continues to Steps 126 and 128, where for each of theparallel streams of interleaved data, the baseband processing modulemaps the interleaved data into a quadrature amplitude modulated (QAM)symbol to produce frequency domain symbols at Step 126. At Step 128, thebaseband processing module converts the frequency domain symbols intotime domain symbols, which may be done utilizing an inverse fast Fouriertransform. The conversion of the frequency domain symbols into the timedomain symbols may further include adding a cyclic prefix to allowremoval of intersymbol interference at the receiver. Note that thelength of the inverse fast Fourier transform and cyclic prefix aredefined in the mode tables of tables 1-12. In general, a 64-pointinverse fast Fourier transform is employed for 20 MHz channels and128-point inverse fast Fourier transform is employed for 40 MHzchannels.

The process then proceeds to Step 130 where the baseband processingmodule space and time encodes the time domain symbols for each of theparallel streams of interleaved data to produce the streams of symbols.In one embodiment, the space and time encoding may be done by space andtime encoding the time domain symbols of the parallel streams ofinterleaved data into a corresponding number of streams of symbolsutilizing an encoding matrix. Alternatively, the space and time encodingmay be done by space and time encoding the time domain symbols ofM-parallel streams of interleaved data into P-streams of symbolsutilizing the encoding matrix, where P=M+1. In one embodiment theencoding matrix may comprise a form of:

$\begin{bmatrix}C_{1} & C_{2} & C_{3} & \ldots & C_{{2M} - 1} \\{- C_{2}^{*}} & C_{1}^{*} & C_{4} & \ldots & C_{2M}\end{bmatrix}\quad$where the number of rows of the encoding matrix corresponds to M and thenumber of columns of the encoding matrix corresponds to P. Theparticular symbol values of the constants within the encoding matrix maybe real or imaginary numbers.

FIG. 7 is a logic diagram of one method that may be utilized by thebaseband processing module to encode the scrambled data at Step 116 ofFIG. 5. In this method, the process begins at Step 140 where thebaseband processing module performs a convolutional encoding with 64state codes and generator polynomials of G₀=133₈ and G₁=171₈ on thescrambled data to produce convolutional encoded data. The process thenproceeds to Step 142 where the baseband processing module punctures theconvolutional encoded data at one of a plurality of rates in accordancewith the mode selection signal to produce the encoded data. Note thatthe puncture rates may include ½, ⅔rds and/or ¾, or any rate asspecified in tables 1-12. Note that, for a particular, mode, the ratemay be selected for backward compatibility with IEEE 802.11(a) and/orIEEE 802.11(g) rate requirements.

The encoding of FIG. 7 may further include an optional Step 144 wherethe baseband processing module combines the convolutional encoding withan outer Reed Solomon code to produce the convolutional encoded data.Note that Step 144 would be conducted in parallel with Step 140.

FIG. 8 is a logic diagram of another encoding method that may beutilized by the baseband processing module to encode the scrambled dataat Step 116 of FIG. 5. In this embodiment, the process begins at Step146 where the baseband processing module encodes the scrambled data inaccordance with a complimentary code keying (CCK) code to produce theencoded data. This may be done in accordance with IEEE 802.11(b)specifications and/or IEEE 802.11(g) specifications. The encoding mayinclude an optional Step 148, which is performed in parallel with Step146 that combines the CCK code with an outer Reed Solomon code toproduce the encoded data.

FIG. 9 is a logic diagram of yet another method for encoding thescrambled data at Step 116, which may be performed by the basebandprocessing module. In this embodiment, the process begins at Step 150where the baseband processing module performs a convolutional encodingwith 256 state codes and generator polynomials of G₀=561₈ and G₁=753₈ onthe scrambled data to produce convolutional encoded data. The processthen proceeds to Step 152 where the baseband processing module puncturesthe convolutional encoded data at one of the plurality of rates inaccordance with a mode selection signal to produce encoded data. Notethat the puncture rate is indicated in the tables 1-12 for thecorresponding mode.

The encoding of FIG. 9 may further include the optional Step 154 wherethe baseband processing module combines the convolutional encoding withan outer Reed Solomon code to produce the convolutional encoded data.

FIGS. 10A and 10B illustrate a schematic block diagram of a multipletransmitter in accordance with the present invention. In FIG. 10A, thebaseband processing is shown to include a scrambler 172, channel encoder174, interleaver 176, demultiplexer 178, a plurality of symbol mappers180-184, a plurality of inverse fast Fourier transform (IFFT)/cyclicprefix addition modules 186-190 and a space/time encoder 192. Thebaseband portion of the transmitter may further include a mode managermodule 175 that receives the mode selection signal 119 and producessettings 121 for the radio transmitter portion and produces the rateselection 117 for the baseband portion.

In operations, the scrambler 172 adds (in GF2) a pseudo random sequenceto the outbound data bits 88 to make the data appear random. A pseudorandom sequence may be generated from a feedback shift register with thegenerator polynomial of S(x)=x⁷+x⁴+1 to produce scrambled data. Thechannel encoder 174 receives the scrambled data and generates a newsequence of bits with redundancy. This will enable improved detection atthe receiver. The channel encoder 174 may operate in one of a pluralityof modes. For example, for backward compatibility with IEEE 802.11(a)and IEEE 802.11(g), the channel encoder has the form of a rate ½convolutional encoder with 64 states and a generator polynomials ofG₀=133₈ and G₁=171₈. The output of the convolutional encoder may bepunctured to rates of ½, ⅔ rds and ¾ according to the specified ratetables (e.g., tables 1-12). For backward compatibility with IEEE802.11(b) and the CCK modes of IEEE 802.11(g), the channel encoder hasthe form of a CCK code as defined in IEEE 802.11(b). For higher datarates (such as those illustrated in tables 6, 8 and 10), the channelencoder may use the same convolution encoding as described above or itmay use a more powerful code, including a convolutional code with morestates, a parallel concatenated (turbo) code and/or a low density paritycheck (LDPC) block code. Further, any one of these codes may be combinedwith an outer Reed Solomon code. Based on a balancing of performance,backward compatibility and low latency, one or more of these codes maybe optimal. Note that the concatenated turbo encoding and low densityparity check will be described in greater detail with reference to FIGS.12-19.

The interleaver 176 receives the encoded data and spreads it overmultiple symbols and transmit streams. This allows improved detectionand error correction capabilities at the receiver. In one embodiment,the interleaver 176 will follow the IEEE 802.11(a) or (g) standard inthe backward compatible modes. For higher performance modes (e.g., suchas those illustrated in tables 6, 8 and 10), the interleaver willinterleave data over multiple transmit streams. The demultiplexer 178converts the serial interleave stream from interleaver 176 intoM-parallel streams for transmission.

Each symbol mapper 180-184 receives a corresponding one of theM-parallel paths of data from the demultiplexer. Each symbol mapper180-182 lock maps bit streams to quadrature amplitude modulated QAMsymbols (e.g., BPSK, QPSK, 16 QAM, 64 QAM, 256 QAM, et cetera) accordingto the rate tables (e.g., tables 1-12). For IEEE 802.11(a) backwardcompatibility, double gray coding may be used.

The map symbols produced by each of the symbol mappers 180-184 areprovided to the IFFT/cyclic prefix addition modules 186-190, whichperforms frequency domain to time domain conversions and adds a prefix,which allows removal of inter-symbol interference at the receiver. Notethat the length of the IFFT and cyclic prefix are defined in the modetables of tables 1-12. In general, a 64-point IFFT will be used for 20MHz channels and 128-point IFFT will be used for 40 MHz channels.

The space/time encoder 192 receives the M-parallel paths of time domainsymbols and converts them into P-output symbols. In one embodiment, thenumber of M-input paths will equal the number of P-output paths. Inanother embodiment, the number of output paths P will equal M+1 paths.For each of the paths, the space/time encoder multiples the inputsymbols with an encoding matrix that has the form of

$\begin{bmatrix}C_{1} & C_{2} & C_{3} & \ldots & C_{{2M} - 1} \\{- C_{2}^{*}} & C_{1}^{*} & C_{4} & \ldots & C_{2M}\end{bmatrix}\quad$Note that the rows of the encoding matrix correspond to the number ofinput paths and the columns correspond to the number of output paths.

FIG. 10B illustrates the radio portion of the transmitter that includesa plurality of digital filter/up-sampling modules 194-198,digital-to-analog conversion modules 200-204, analog filters 206-216,I/Q modulators 218-222, RF amplifiers 224-228, RF filters 230-234 andantennas 236-240. The P-outputs from the space/time encoder 192 arereceived by respective digital filtering/up-sampling modules 194-198.

In operation, the number of radio paths that are active correspond tothe number of P-outputs. For example, if only one P-output path isgenerated, only one of the radio transmitter paths will be active. Asone of average skill in the art will appreciate, the number of outputpaths may range from one to any desired number.

The digital filtering/up-sampling modules 194-198 filter thecorresponding symbols and adjust the sampling rates to correspond withthe desired sampling rates of the digital-to-analog conversion modules200-204. The digital-to-analog conversion modules 200-204 convert thedigital filtered and up-sampled signals into corresponding in-phase andquadrature analog signals. The analog filters 208-214 filter thecorresponding in-phase and/or quadrature components of the analogsignals, and provide the filtered signals to the corresponding I/Qmodulators 218-222. The I/Q modulators 218-222 based on a localoscillation, which is produced by a local oscillator 100, up-convertsthe I/Q signals into radio frequency signals.

The RF amplifiers 224-228 amplify the RF signals which are thensubsequently filtered via RF filters 230-234 before being transmittedvia antennas 236-240.

FIGS. 11A and 11B illustrate a schematic block diagram of anotherembodiment of a receiver in accordance with the present invention. FIG.11A illustrates the analog portion of the receiver which includes aplurality of receiver paths. Each receiver path includes an antenna, RFfilters 252-256, low noise amplifiers 258-260, I/Q demodulators 264-268,analog filters 270-280, analog-to-digital converters 282-286 and digitalfilters and down-sampling modules 288-290.

In operation, the antennas receive inbound RF signals, which areband-pass filtered via the RF filters 252-256. The corresponding lownoise amplifiers 258-260 amplify the filtered signals and provide themto the corresponding I/Q demodulators 264-268. The I/Q demodulators264-268, based on a local oscillation, which is produced by localoscillator 100, down-converts the RF signals into baseband in-phase andquadrature analog signals.

The corresponding analog filters 270-280 filter the in-phase andquadrature analog components, respectively. The analog-to-digitalconverters 282-286 convert the in-phase and quadrature analog signalsinto a digital signal. The digital filtering and down-sampling modules288-290 filter the digital signals and adjust the sampling rate tocorrespond to the rate of the baseband processing, which will bedescribed in FIG. 11B.

FIG. 11B illustrates the baseband processing of a receiver. The basebandprocessing includes a space/time decoder 294, a plurality of fastFourier transform (FFT)/cyclic prefix removal modules 296-300, aplurality of symbol demapping modules 302-306, a multiplexer 308, adeinterleaver 310, a channel decoder 312, and a descramble module 314.The baseband processing module may further include a mode managingmodule 175. The space/time decoding module 294, which performs theinverse function of space/time encoder 192, receives P-inputs from thereceiver paths and produce M-output paths. The M-output paths areprocessed via the FFT/cyclic prefix removal modules 296-300 whichperform the inverse function of the IFFT/cyclic prefix addition modules186-190 to produce frequency domain symbols.

The symbol demapping modules 302-306 convert the frequency domainsymbols into data utilizing an inverse process of the symbol mappers180-184. The multiplexer 308 combines the demapped symbol streams into asingle path.

The deinterleaver 310 deinterleaves the single path utilizing an inversefunction of the function performed by interleaver 176. The deinterleaveddata is then provided to the channel decoder 312 which performs theinverse function of channel encoder 174. The descrambler 314 receivesthe decoded data and performs the inverse function of scrambler 172 toproduce the inbound data 98. The baseband portion of the transmitter mayfurther include a mode manager module 175 that receives the modeselection signal 119 and produces settings 121 for the radio receiverportion and produces the rate selection 117 for the baseband portion.

FIG. 12 is a schematic block diagram of channel encoder 174 implementedas a turbo encoder. In this embodiment, the turbo encoder receives inputbits, modifies them 321, processes them via a constituent encoder320-322 and interleaves them to produce the corresponding encodedoutput. Depending on the particular symbol mapping (BPSK, QPSK, 8PSK(phase shift keying), 64 QAM, 16 QAM or 16APSK (amplitude phase shiftkeying), the turbo encoder will function in the same manner to producethe encoded data. For instance, of π₀ and π₁ are interleaves of MSB(most significant bit) and LSB (least significant bit), respectively,for a block of 2-bit symbols and π_(L) ⁻¹, L=0, are the inverses, thenthe modified interleave is as follows:

${{\pi^{''}{l(i)}} = {\begin{pmatrix}{{i\text{:}i\;{mod}\; 2} = 0} \\{{\pi\; - {1(i)\text{:}i\;{mod}\; 2}} = 1}\end{pmatrix}\mspace{14mu}{and}}}\mspace{14mu}$${\pi\;{l(i)}} = \begin{pmatrix}{{i\text{:}i\;{mod}\; 2} = 1} \\{{{\pi(i)}\text{:}i\;{mod}\; 2} = 0}\end{pmatrix}$

FIG. 13 illustrates an embodiment of the constituent encoders 320-322 ofFIG. 12 which may be implemented as rate ½ encoders. As shown, theencoder 320-322 includes summing modules 325, 329, 335, 337 and digitalone-cycle delays 327, 331, 333.

FIG. 14 illustrates a schematic block diagram of another embodiment of aconstituent encoder 320-322 that utilizes the ½ rate encoder 345 toproduce a rate ⅖ths encoder. In this embodiment, two consecutive binaryinputs 341, 434 are sent to the rate ½ encoder 345 and to a summingmodule 339. The output of the rate ⅖ encoder is produced as shown.

FIG. 15 represents the generally functionality of FIG. 14. The rate ⅖thsencoder 347 may then be utilized as puncture encoders 347 as shown inFIGS. 16 and 17, which have the corresponding QPSK mapping.

FIG. 18 illustrates the channel encoder 174 being implemented as a lowdensity parity check (LDPC) encoder. In this embodiment, the encoderincludes a low density parity check encoder 174, an interleaver 176 anda gray mapping module 177. The block length may be 2000 and theinformation length may be 1600. In this instance, the low density paritycheck binary matrix H=[H₁, H₂], where H₁ is an irregular 400×1600 lowdensity matrix with 1400 columns of weight 3 and 200 columns of weight7, and all rows of weight 14. More over, the distribution of the 1's ispseudo random in order to suit a hardware embodiment. The matrix H₂ is a400×400 matrix which provides a long path with no loops in the bipartitegraph between redundancy bit node and check node.

${H\; 2} = {\begin{matrix}{100\mspace{14mu}\ldots\mspace{14mu} 00} \\{11\_\mspace{14mu}\ldots\mspace{14mu} 00} \\{\_ 11\mspace{14mu}\ldots\mspace{14mu} 00} \\\ldots \\{000\mspace{14mu}\ldots\mspace{14mu} 10} \\{000\mspace{14mu}\ldots\mspace{14mu} 11}\end{matrix}}$This parity check matrix provides easy encoding. The code has no circleof loops less than 6. The degree distribution of the bipartite graph ofthe code is listed in the following table. The total number of edges ofthe graph is 6399.

number of nodes bit node degree (number of edges emitted from a bitnode) 1 1 2 399 3 1400 7 200 check node degree 15 1 16 399

FIG. 19 illustrates a particular interleaving 351, 353 and QPSK map 355that may be utilized by the encoder of FIG. 18. In this embodiment, therate of the code may be ½ and the LDPC code is symmetric. As such, theinterleaving is as shown.

FIG. 20 is a diagram of an example asymmetric communication between anaccess point (AP) 361 and a wireless communication device (e.g., apersonal computer (PC)) 363. In this example, there are 4 transmittingantennas at the AP 361, which may be represented by M, and two receivingantennas at the PC 363, which may be represented by N. In this example,M>N.

FIG. 21 is a diagram of space-time block coding that plots time versusfrequency versus space (i.e., antennas). As shown, the frequency isdividing into a plurality of subcarriers (e.g., 64 for a 20 MHz channel,128 for a 40 MHz channel). As is also shown, the space is divided into anumber of antennas. In this illustration, only two antennas are shown,but more may be used. As is further shown, time is divided into aplurality of symbols. Thus, for each time interval, each subcarrier oneach antenna supports a particular symbol. The space time block codingwill be further described with reference to FIGS. 22-26.

FIG. 22 is a diagram of two antenna asymmetrical wireless communication.Such a communication may be implemented using simple spatialmultiplexing where different information is transmitted on each of thetwo antennas. Alternatively, space time block coding may be used. Inthis instance, with a rate=1 Alamouti code, the symbol allocation is asshown, where s0 is the first constellation point transmitted onsubcarrier k and s1 is the second constellation point transmitted onsubcarrier k+1.

FIG. 23 is a diagram of three antenna asymmetrical wirelesscommunication. In this example, on each OFDM subcarrier index k, theencoded transmission is as shown, where the * indicates a complexconjugate. With this configuration, one rate=1 Alamouti space time blockcode (STBC) stream may be transmitted along with one uncoded stream perOFMD subcarrier k with a coding delay of two symbol periods.

FIG. 24 is another diagram of three antenna asymmetrical wirelesscommunication. This is like the communication of FIG. 23, except thatthe STBC is applied to alternating pairs of antennas. The decoding delayis still two symbol periods. As shown, the STBC is across antennas {0,1} in the first two symbol periods, then {0, 2} in the next two symbolperiods, then {2, 3} in the next two. Note that, with the channel code(e.g. constraint length=6 convolutional code) across bits, this mightimprove the robustness of the lowest rates.

FIG. 25 is a diagram of four antenna asymmetrical wirelesscommunication. In this diagram, some sort of coding is implemented toensure reliable transmission to, at a minimum, an N=2 station. Further,a transmit two rate=1 Alamouti space-time block coded streams per OFDMsubcarrier index k may be constructed as shown. The decoding delay istwo symbol periods.

FIG. 26 is another diagram of four antenna asymmetrical wirelesscommunication. This is similar to the communication of FIG. 25, exceptthat the STBC is applied to alternating pairs of antennas. The decodingdelay is still two symbol periods. For example, the STBC is acrossantennas {0, 1} and {2, 3} in the first two symbol periods, then {0, 2}and {1, 3}. Note that, with the channel code (e.g. constraint length=6convolutional code) across bits, this might improve the robustness ofthe lowest rates.

As one of average skill in the art will appreciate, the term“substantially” or “approximately”, as may be used herein, provides anindustry-accepted tolerance to its corresponding term. Such anindustry-accepted tolerance ranges from less than one percent to twentypercent and corresponds to, but is not limited to, component values,integrated circuit process variations, temperature variations, rise andfall times, and/or thermal noise. As one of average skill in the artwill further appreciate, the term “operably coupled”, as may be usedherein, includes direct coupling and indirect coupling via anothercomponent, element, circuit, or module where, for indirect coupling, theintervening component, element, circuit, or module does not modify theinformation of a signal but may adjust its current level, voltage level,and/or power level. As one of average skill in the art will alsoappreciate, inferred coupling (i.e., where one element is coupled toanother element by inference) includes direct and indirect couplingbetween two elements in the same manner as “operably coupled”. As one ofaverage skill in the art will further appreciate, the term “comparesfavorably”, as may be used herein, indicates that a comparison betweentwo or more elements, items, signals, etc., provides a desiredrelationship. For example, when the desired relationship is that signal1 has a greater magnitude than signal 2, a favorable comparison may beachieved when the magnitude of signal 1 is greater than that of signal 2or when the magnitude of signal 2 is less than that of signal 1.

The preceding discussion has presented various embodiments of a multipleinput/multiple output transceiver for use in wireless communicationsystems. As one of average skill in the art will appreciate, otherembodiments may be derived from the teaching of the present inventionwithout deviating from the scope of the claims.

Mode Selection Tables:

TABLE 1 2.4 GHz, 20/22 MHz channel BW, 54 Mbps max bit rate Code RateModulation Rate NBPSC NCBPS NDBPS EVM Sensitivity ACR AACR Barker 1 BPSKBarker 2 QPSK 5.5 CCK 6 BPSK 0.5 1 48 24 −5 −82 16 32 9 BPSK 0.75 1 4836 −8 −81 15 31 11 CCK 12 QPSK 0.5 2 96 48 −10 −79 13 29 18 QPSK 0.75 296 72 −13 −77 11 27 24 16-QAM 0.5 4 192 96 −16 −74 8 24 36 16-QAM 0.75 4192 144 −19 −70 4 20 48 64-QAM 0.666 6 288 192 −22 −66 0 16 54 64-QAM0.75 6 288 216 −25 −65 −1 15

TABLE 2 Channelization for Table 1 Channel Frequency (MHz) 1 2412 2 24173 2422 4 2427 5 2432 6 2437 7 2442 8 2447 9 2452 10 2457 11 2462 12 2467

TABLE 3 Power Spectral Density (PSD) Mask for Table 1 PSD Mask 1Frequency Offset dBr −9 MHz to 9 MHz 0 +/−11 MHz −20 +/−20 MHz −28 +/−30MHz and greater −50

TABLE 4 5 GHz, 20 MHz channel BW, 54 Mbps max bit rate Code RateModulation Rate NBPSC NCBPS NDBPS EVM Sensitivity ACR AACR 6 BPSK 0.5 148 24 −5 −82 16 32 9 BPSK 0.75 1 48 36 −8 −81 15 31 12 QPSK 0.5 2 96 48−10 −79 13 29 18 QPSK 0.75 2 96 72 −13 −77 11 27 24 16-QAM 0.5 4 192 96−16 −74 8 24 36 16-QAM 0.75 4 192 144 −19 −70 4 20 48 64-QAM 0.666 6 288192 −22 −66 0 16 54 64-QAM 0.75 6 288 216 −25 −65 −1 15

TABLE 5 Channelization for Table 4 Frequency Channel Frequency (MHz)Country Channel (MHz) Country 240 4920 Japan 244 4940 Japan 248 4960Japan 252 4980 Japan 8 5040 Japan 12 5060 Japan 16 5080 Japan 36 5180USA/Europe 34 5170 Japan 40 5200 USA/Europe 38 5190 Japan 44 5220USA/Europe 42 5210 Japan 48 5240 USA/Europe 46 5230 Japan 52 5260USA/Europe 56 5280 USA/Europe 60 5300 USA/Europe 64 5320 USA/Europe 1005500 USA/Europe 104 5520 USA/Europe 108 5540 USA/Europe 112 5560USA/Europe 116 5580 USA/Europe 120 5600 USA/Europe 124 5620 USA/Europe128 5640 USA/Europe 132 5660 USA/Europe 136 5680 USA/Europe 140 5700USA/Europe 149 5745 USA 153 5765 USA 157 5785 USA 161 5805 USA 165 5825USA

TABLE 6 2.4 GHz, 20 MHz channel BW, 192 Mbps max bit rate ST TX CodeModu- Code Rate Antennas Rate lation Rate NBPSC NCBPS NDBPS 12 2 1 BPSK0.5 1 48 24 24 2 1 QPSK 0.5 2 96 48 48 2 1 16-QAM 0.5 4 192 96 96 2 164-QAM 0.666 6 288 192 108 2 1 64-QAM 0.75 6 288 216 18 3 1 BPSK 0.5 148 24 36 3 1 QPSK 0.5 2 96 48 72 3 1 16-QAM 0.5 4 192 96 144 3 1 64-QAM0.666 6 288 192 162 3 1 64-QAM 0.75 6 288 216 24 4 1 BPSK 0.5 1 48 24 484 1 QPSK 0.5 2 96 48 96 4 1 16-QAM 0.5 4 192 96 192 4 1 64-QAM 0.666 6288 192 216 4 1 64-QAM 0.75 6 288 216

TABLE 7 Channelization for Table 6 Channel Frequency (MHz) 1 2412 2 24173 2422 4 2427 5 2432 6 2437 7 2442 8 2447 9 2452 10 2457 11 2462 12 2467

TABLE 8 5 GHz, 20 MHz channel BW, 192 Mbps max bit rate ST TX Code Modu-Code Rate Antennas Rate lation Rate NBPSC NCBPS NDBPS 12 2 1 BPSK 0.5 148 24 24 2 1 QPSK 0.5 2 96 48 48 2 1 16-QAM 0.5 4 192 96 96 2 1 64-QAM0.666 6 288 192 108 2 1 64-QAM 0.75 6 288 216 18 3 1 BPSK 0.5 1 48 24 363 1 QPSK 0.5 2 96 48 72 3 1 16-QAM 0.5 4 192 96 144 3 1 64-QAM 0.666 6288 192 162 3 1 64-QAM 0.75 6 288 216 24 4 1 BPSK 0.5 1 48 24 48 4 1QPSK 0.5 2 96 48 96 4 1 16-QAM 0.5 4 192 96 192 4 1 64-QAM 0.666 6 288192 216 4 1 64-QAM 0.75 6 288 216

TABLE 9 channelization for Table 8 Frequency Channel Frequency (MHz)Country Channel (MHz) Country 240 4920 Japan 244 4940 Japan 248 4960Japan 252 4980 Japan 8 5040 Japan 12 5060 Japan 16 5080 Japan 36 5180USA/Europe 34 5170 Japan 40 5200 USA/Europe 38 5190 Japan 44 5220USA/Europe 42 5210 Japan 48 5240 USA/Europe 46 5230 Japan 52 5260USA/Europe 56 5280 USA/Europe 60 5300 USA/Europe 64 5320 USA/Europe 1005500 USA/Europe 104 5520 USA/Europe 108 5540 USA/Europe 112 5560USA/Europe 116 5580 USA/Europe 120 5600 USA/Europe 124 5620 USA/Europe128 5640 USA/Europe 132 5660 USA/Europe 136 5680 USA/Europe 140 5700USA/Europe 149 5745 USA 153 5765 USA 157 5785 USA 161 5805 USA 165 5825USA

TABLE 10 5 GHz, with 40 MHz channels and max bit rate of 486 Mbps TX STCode Code Rate Antennas Rate Modulation Rate NBPSC 13.5 Mbps  1 1 BPSK0.5 1  27 Mbps 1 1 QPSK 0.5 2  54 Mbps 1 1 16-QAM 0.5 4 108 Mbps 1 164-QAM 0.666 6 121.5 Mbps   1 1 64-QAM 0.75 6  27 Mbps 2 1 BPSK 0.5 1 54 Mbps 2 1 QPSK 0.5 2 108 Mbps 2 1 16-QAM 0.5 4 216 Mbps 2 1 64-QAM0.666 6 243 Mbps 2 1 64-QAM 0.75 6 40.5 Mbps  3 1 BPSK 0.5 1  81 Mbps 31 QPSK 0.5 2 162 Mbps 3 1 16-QAM 0.5 4 324 Mbps 3 1 64-QAM 0.666 6 365.5Mbps   3 1 64-QAM 0.75 6  54 Mbps 4 1 BPSK 0.5 1 108 Mbps 4 1 QPSK 0.5 2216 Mbps 4 1 16-QAM 0.5 4 432 Mbps 4 1 64-QAM 0.666 6 486 Mbps 4 164-QAM 0.75 6

TABLE 11 Power Spectral Density (PSD) mask for Table 10 PSD Mask 2Frequency Offset dBr −19 MHz to 19 MHz 0 +/−21 MHz −20 +/−30 MHz −28+/−40 MHz and greater −50

TABLE 12 Channelization for Table 10 Frequency Channel Frequency (MHz)Country Channel (MHz) County 242 4930 Japan 250 4970 Japan 12 5060 Japan38 5190 USA/Europe 36 5180 Japan 46 5230 USA/Europe 44 5520 Japan 545270 USA/Europe 62 5310 USA/Europe 102 5510 USA/Europe 110 5550USA/Europe 118 5590 USA/Europe 126 5630 USA/Europe 134 5670 USA/Europe151 5755 USA 159 5795 USA

As an example of gains that can be obtained from MIMO, channelmeasurements may be used. In this example, assume K sub-channel OFDMsystem with equal transmit power per tone. The equal power (i.e. nospace-time waterfilling) N×N MIMO capacity per sub-channel is [Telatar,Foschini and Gans]:

$C_{k} = {\log_{2}\left( {\det\left\lbrack {I_{N} + {\frac{\rho}{N}H_{k}^{H}H_{k}}} \right\rbrack} \right)}$Where k—tone number, K—number of tones, Hk—N×N channel matrix (pertone), N—number of antennas, ρ=SNR/κ=normalized receiver SNR, and

$\kappa = {\frac{1}{{KN}^{2}}{\sum\limits_{k = 0}^{K - 1}{\sum\limits_{m = 0}^{N - 1}{\sum\limits_{n = 0}^{N - 1}{{H_{k}^{*}\left( {n,m} \right)}{H_{k}\left( {n,m} \right)}}}}}}$Accordingly, singular values of the channel matrix for each sub-channeldetermine the capacity of that sub-channel, where the singular valuedecomposition may be defined as

$H_{k} = {{U_{k}\Sigma_{k}V_{k}^{H}\mspace{14mu}{with}\mspace{14mu}\Sigma_{k}} = \begin{bmatrix}\sigma_{1k} & \; & \; \\\; & \cdot & \; \\\; & \; & \sigma_{Nk}\end{bmatrix}}$Then capacity of kth subchannel can then be expressed in terms of

$C_{k} = {\sum\limits_{n = 0}^{N - 1}{\log_{2}\left( {1 + {\frac{\rho}{N}\sigma_{n,k}^{2}}} \right)}}$As can be seen, capacity is maximized when the singular values are ofequal value.

From this example, expected MIMO performance gains include: (a) undermoderate SNR conditions, MIMO can achieve significant data rateimprovements, where the data rate increase is proportional to N (# of TXantennas) and will require higher SNR. Note that the range reduction maybecome more severe as N increases. (b) N should be selected based onpractical considerations such as, but not limited to, implementationcomplexity and maximum target rate that is competitive with othertechniques. In one embodiment, max N=4.

1. A method for asymmetrical MIMO wireless communication, the method comprises: determining a number of transmission antennas for the asymmetrical MIMO wireless communication; determining a number of reception antennas for the asymmetrical MIMO wireless communication; determining whether the asymmetric MIMO wireless communication will use spatial multiplexing or space-time block coding by: using the spatial time block coding whenever the number of reception antennas is less than the number of transmission antennas, wherein, when both, the number of reception antennas is an even number and the number of transmission antennas exceeds the number of reception antennas by an odd number, the spatial time block coding comprises: selecting one of the transmission antennas for transmitting non-spatial time block coding constellation points during first and second time intervals.
 2. The method of claim 1 further comprising: using the spatial multiplexing whenever the number of reception antennas equals the number of transmission antennas.
 3. The method of claim 1, wherein the determining the number of reception antennas comprises: receiving, from a receiving wireless communication device, the number of reception antennas.
 4. The method of claim 1, wherein the spatial time block coding further comprises: grouping remaining transmission antennas of the transmission antennas into at least one group of transmission antennas.
 5. The method of claim 4, wherein the at least one group of transmission antennas transmits spatial time block coding constellation points.
 6. The method of claim 1, wherein the selecting one of the transmission antennas comprises: from spatial-time coding interval to spatial-time coding interval, selecting another one of the transmission antennas for transmitting the non-spatial time block coding constellation points.
 7. The method of claim 1, wherein, when the number of transmission antennas exceeds the number of reception antennas by an even number, the spatial time block coding comprises: dividing the transmission antennas into at least two groups, where each of the at least two groups includes at least two of the transmission antennas; and individually performing the spatial time block coding for the at least two groups.
 8. The method of claim 7 further comprises: from spatial-time coding interval to spatial-time coding interval, dividing the transmission antennas into at least two different groups.
 9. A method for asymmetrical MIMO wireless communication, the method comprises: determining a number of transmission antennas for the asymmetrical MIMO wireless communication; determining a number of reception antennas for the asymmetrical MIMO wireless communication; whenever the number of transmission antennas exceeds the number of reception antennas, using spatial time block coding for the asymmetrical MIMO wireless communication, wherein, when both the number of reception antennas is an even number and the number of transmission antennas exceeds the number of reception antennas by an odd number, the spatial time block coding comprises: selecting one of the transmission antennas for transmitting non-spatial time block coding constellation points during first and second time intervals.
 10. The method of claim 9, wherein the determining the number of reception antennas comprises: receiving, from a receiving wireless communication device, the number of reception antennas.
 11. The method of claim 9, further comprising: whenever the number of transmission antennas does not exceed the number of reception antennas, using spatial multiplexing for the asymmetrical MIMO wireless communication.
 12. The method of claim 9, wherein the selecting one of the transmission antennas comprises: from spatial-time coding interval to spatial-time coding interval, selecting another one of the transmission antennas for transmitting the non-spatial time block coding constellation points.
 13. The method of claim 9, wherein, when the number of transmission antennas exceeds the number of reception antennas by an even number, the spatial time block coding comprises: dividing the transmission antennas into at least two groups, where each of the at least two groups includes at least two of the transmission antennas; and individually performing the spatial time block coding for the at least two groups.
 14. The method of claim 13 further comprises: from spatial-time coding interval to spatial-time coding interval, dividing the transmission antennas into at least two different groups.
 15. The method of claim 9, wherein the spatial time block coding further comprises: grouping remaining transmission antennas of the transmission antennas into at least one group of transmission antennas.
 16. The method of claim 15, wherein the at least one group of transmission antennas transmits spatial time block coding constellation points. 